In a conventional telecommunication system employing frequency hopping techniques a transmitter transmits an information waveform, such as a voice signal or a digital signal, over a transmission medium at various transmission or channel frequencies. For instance, the transmitter transmits the information waveform at a first channel frequency for a first period of time, transmits at a second channel frequency for a second period of time, etc. In other words, the channel frequency remains constant for a period of time and then changes or hops to a new channel frequency. The rate at which a transmitter hops to a new frequency is called the hopping rate. A frequency hopping transmitter transmits the information waveform at channel frequencies determined according to a sequence of channel frequencies, called a frequency hopping sequence, and the hopping rate. A frequency hopping algorithm determines the sequence of channel frequencies in the frequency hopping sequence. A simple form of the frequency hopping algorithm consists of a predetermined sequence of channel frequencies. Alternatively, the frequency hopping algorithm consists of a complex equation which generates the next channel frequency.
A receiver receives the information waveform via the transmission medium, and outputs the information waveform. The receiver receives and demodulates the information in accordance with the same frequency hopping algorithm employed by the transmitter. Therefore, the frequency hopping receiver hops to a new channel frequency in synchronization with the transmitter.
Conventional transmitters and receivers employ a crystal standard to generate a standard frequency. A desired channel frequency is then derived from the standard frequency according to well known techniques. Crystal standards have the drawback that the standard frequency generated changes with changes in temperature. This causes frequency error in transmitters and in receivers using crystal standards to derive the channel frequencies. Typically, the frequency error is linearly related or proportional to the channel frequency. The frequency error, however, may be non-linearly related to the channel frequency.
The total frequency error across a link between transmitter and receiver is the sum of the transmitter frequency error and receiver frequency error. In general, the transmitter frequency error will be different from the receiver frequency error. Therefore, a transmitter with a transmitter frequency error of -3 Hz and a receiver with a receiver frequency error of 8 Hz at a given channel frequency will have a total frequency error of 5 Hz due to a reflection around the base band.
A conventional solution to the above discussed problems is to employ crystal standards of such high quality that the frequency errors generated thereby are negligible. The costs of such systems, however, unacceptably limits their feasibility as a solution.
Another conventional solution is the use of crystal ovens. Crystal ovens are a means of maintaining the crystal standard at a constant temperature. These ovens, however, require a significant amount of time to heat up and consume power, thus, restricting their use in several applications. Furthermore, crystal ovens add to the expense of a telecommunications system, and the added weight of crystal ovens makes their use impractical with many types of mobile transmitters and receivers.
While presenting an economical and viable solution to the problems discussed above, the present invention further contemplates a linked compression-expansion (Lincompex) telecommunications system employing frequency hopping techniques. Lincompex systems are well known in the telecommunications art. Such a digital Lincompex system is taught in U.S. Pat. No. 4,271,499 (the "'499 patent") issued Jun. 2, 1981, to Leveque, the Inventor of the present application. The '499 patent only teaches the transmitting of a voice signal over a Lincompex system. However, it has also been found that Lincompex systems can be utilized to transmit data having a complex waveform. Such a system that overcomes the inherent problem of data and envelope overlap (i.e., complex waveform) is fully disclosed in U.S. Pat. No. 4,907,217 (the "'217 patent"), also to Leveque, issued on Mar. 6, 1990, entitled "System and Method of Transmitting A Complex Waveform Over a Communication Channel Utilizing Lincompex Techniques".
In the voice transmission system, as disclosed in the '499 patent, a voice signal is transmitted using Lincompex techniques. This voice signal is band limited and exhibits a control tone 4 (according to the teachings of the '499 patent) which is also band limited and does not overlap the voice band 2, (see FIG. 2(a)).
This system, however, has not been able to compensate for frequency drift or the detuning of the transmitter/receiver system. It is necessary to synchronize the center frequency of the transmitter and receiver to ensure that the control tone and voice signal are not distorted upon reception.
During transmission of the voice signal, frequency drift or the detuning of the transmitter/receiver system can cause a communication problem with the reproduction of the voice signal. More specifically, the control tone used in the reproduction of the compressed voice signal is very sensitive to a change in the frequency, frequency drift. If the frequency of the control tone drifts due to various factors such as detuning of the transmitter/receiver system, (frequency drift of transmitter or receiver modulation oscillators), the power in the reproduced voice becomes weaker in proportion to this frequency change. In the typical Lincompex System, the relationship of the attenuation in the reproduced voice signal and the change in frequency is governed by the ratio 1 db/2 hz. For example, a 30 hz change or drift in frequency can cause a 15 db error in attenuation in the reproduced voice signal. A system for resolving this problem is fully disclosed in U.S. Pat. No. 5,065,451 (the "'451 patent")issued on Nov. 12, 1991, to Leveque, the Inventor of the present invention. FIGS. 1(a) and 3 illustrates the Lincompex System of the '451 patent for overcoming the above discussed problem.
In the modulator of FIG. 1(a) an input 20 of the Lincompex system modulator receives voice information for transmission. A control tone generator or envelope circuit 24 monitors the input voice signal. A compressor 22 compresses the introduced input voice signal. Compression is performed by dividing the signal by its envelope in pseudo-real time to produce a compressed voice signal. To develop the control tone, the system of FIG. 1(a) supplies the envelope signal developed from the output of the envelope detector 26 to a logarithmic (log) amplifier 28 which then develops a signal representative of the logarithm of the envelope signal. The output of this logarithmic amplifier 28 is supplied to a control terminal of a voltage control FM oscillator 30, which generates a frequency that varies about a center frequency F.sub.c in relation to the variation of the input voltage supplied to its control terminal from the logarithmic amplifier 28, to develop an envelope signal as an output of the control tone generator (envelope circuit) 24.
A summer 32 then sums the compressed voice signal developed at the output of the compressor 22 with the envelope signal developed at the output of the FM oscillator 30 to form a combined information signal.
In this speech transmission system, this combined information signal output from the summer 32 is provided to a transmitter 34 which transmits the signal over a desired transmission medium 36. In a typical embodiment, a single side band transmitter would normally transmit the modulated combined information signal across the airways in a known manner.
As shown in FIG. 3, the demodulator of the Lincompex System for the '451 patent utilizes a frequency compensation circuit which corrects the frequency of the incoming combined information waveform such that the attenuation problem is substantially eliminated.
As illustrated in FIG. 3, the prior art Lincompex demodulator includes a receiving unit 15 which receives a modulated combined information signal from the transmission medium 36, which normally includes an antenna for receiving radio waves from the atmosphere, which supplies the received modulated combined information signal to a receiver 38. The receiver 38 demodulates the transmitted signal to produce a demodulated version of the combined information signal. Typically, this receiver would be a single side band receiver which mixes the received modulated combined information signal with the channel frequency to produce a base band combined information signal.
To recover only the voice signal from such a combined voice/envelope information signal, a low pass filter 40 removes the voice information 2 of FIG. 2(a) from the combined information signal to recover the compressed voice signal containing only the voice information 2. This compressed voice signal is transmitted according to the Lincompex techniques at a substantially complete modulation of the transmitter 34 of FIG. 1(a). The compressed voice signal is passed through a fading regulator 58. The fading regulator 58 is a fading regulator similar to the one described in the '499 patent which removes any audio level variations not removed by the automatic gain control circuitry in the receiver 38. This compressed modified voice signal output from the fading regulator 58 is then expanded to produce the necessary dynamic range for the recovered voice signal to be supplied at the output 60.
Accordingly, an expander 42, similar to the one described in the '499 patent, is utilized which essentially multiplies the compressed modified voice signal developed at the output of the fading regulator 58 by an envelope signal which is recovered by a control tone conversion circuit 27.
The control tone conversion circuit 27 comprises a band pass filter 46 which recovers only the control tone which is represented by the control tone information signal 4 in FIG. 2(a). The control tone originally developed by the FM oscillator 30 of FIG. 1(a) is then passed through a frequency discriminator 48. The frequency discriminator 48 measures the instantaneous frequency of the control tone and produces a voltage level representative of this measured frequency. In this case, a the voltage level also represents the logarithm of the original envelope signal. The antilogarithm circuit 50 is utilized to recover the original envelope signal. This logarithmic signal is then supplied to an anti-logarithm circuit 50. The antilogarithm circuit 50 is utilized to recover the original envelope developed by the envelope detector 26 of FIG. 1(a). This original envelope signal is used by the expander 42 to recover the original voice signal by expanding the compressed voice signal to provide the original signal to the output 60. This control tone conversion circuit 27 is similar to the control tone conversion circuit described in the '499 patent.
The control tone conversion circuit 27 also comprises a calibration tone detector circuit 52 and a frequency measuring circuit 54. The calibration tone detector circuit 52 detects the initial calibration tone transmitted by the transmitter to determine when the frequency of the calibration tone is to be compared with a reference frequency to determine a frequency error.
This calibration tone is generated in a manner similar to the process disclosed in the '499 patent. A control unit 88 outputs control signals to the control tone generator 24, switch 86 and switch 87. As discussed in the '499 patent, the control unit 88 is a transmit key detect circuit which detects transmitter key-down. In response to the control signals, the calibration tone generator 24 outputs an unattenuated and unmodulated signal of a predetermined frequency, and the switch 87 disconnects the compressor 22 from the summer 32. Thus, the summer 32 only outputs the unattenuated and unmodulated signal of predetermined frequency. The unattenuated and unmodulated signal of predetermined frequency can serve as the calibration tone, or the switch 86, in response to the control signals, disconnects and reconnects the summer 32 to the transmitter 34 to create a predetermined pattern of the unattenuated and unmodulated signal as the calibration tone. Alternatively, switch 86 is on attenuator which, in response to the control signals, disables and enables attenuation of the unmodulated and unattenuated signal according to a predetermined pattern. Furthermore, the attenuator could switch between anti-attenuation and attenuation. For instance, the anti-attenuation could increase by 5 db the unmodulated and unattenuated signal, and then attenuate by 31 db the unmodulated and unattenuated signal.
The predetermined frequency of the calibration tone preferably corresponds to the center frequency of voltage control FM oscillator 30, but is not limited thereto. Furthermore, the control unit 88 preferably causes the calibration tone to be generated for a 280-300 millisecond duration, but is not limited thereto. After sending the calibration tone for a given duration, the control unit 88 causes switches 86 and 87 to close, and permits the control tone generator 24 to output the control tone.
As a further alternative, the calibration tone can be generated as shown in FIG. 1(b). In FIG. 1(b), the switch 86, in response to a control signal from calibration tone generator 89, disconnects the transmitter 34 from the summer 32 and connects the transmitter 34 to the calibration tone generator 89. The calibration tone generator 89 can include a function generator (not shown) which generates the calibration tone for transmission by transmitter 34.
The calibration tone detector circuit 52 generates a control signal representing that a calibration tone has been received. The detection of the calibration tone utilizes the conventional method described in the '499 patent. The frequency measuring circuit 54 receives this signal from the calibration tone detector circuit 52 and compares this signal representative of the frequency of the calibration tone as generated by the discriminator 48 with the desired standard frequency, i.e, the intended control tone center frequency. If a difference in the frequencies is determined, the frequency measuring circuit 54 recognizes that frequency error is present and generates a frequency error signal, which is stored and used by a frequency compensation circuit 56 so that the frequency compensation circuit 56 can correctly frequency translate the frequency of the combined information waveform. The above frequency error determination can be implemented using software techniques.
An example of the frequency compensation circuit 56 is a typical phase shifting circuit or frequency translator described in the '451 patent and in FIG. 6(a). These circuits usually include a Hilbert Transform Circuit, an oscillator, phase shifter, multipliers, and a summer. In this example, (as shown in FIG. 6(a)), the signal path is split into two paths, A and B. Path A is connected to a multiplier 203. Connected to multiplier 203 is a voltage controlled oscillator 205 and a summer 211. The voltage controlled oscillator 205 is also connected to a 90.degree. phase shifting device 209. Furthermore, Path B is connected to a Hilbert Transform Circuit 201. Connected to the Hilbert Transform Circuit 201 is a multiplier 207. This multiplier 207 is connected to the 90.degree. phase shifting device 209 and the summer 211. The summer is connected to the input of the Lincompex demodulator.
An incoming signal is split into two paths, A and B. The signal traveling along path A is modulated, multiplied, by multiplier 203, with a cosine waveform type signal generated by the voltage controlled oscillator 205. The frequency of this cosine signal is determined according to the frequency error measured in the frequency measuring circuit 54 with a range between 0 to 100 Hz. The modulated signal from multiplier 203 is supplied to summer 211 to be added with a signal modified along path B.
The signal traveling along path B is first passed through a Hilbert Transform Circuit 201 which shifts the positive frequency components of the signal by -90.degree. and shifts the negative frequency components of the signal by +90.degree.. After being transformed by the Hilbert Transform Circuit 201, the signal is modulated, multiplied, by multiplier 207, with a sine waveform type signal. This sine signal is a 90.degree. shifted version of the cosine signal generated by the voltage controlled oscillator 205. The frequency of this signal is equal to the frequency of the cosine signal. The modulated signal is supplied to the summer 211 so that its negative and positive components can be summed with the modulated signal of path A. The signal generated by the summer 214 is a frequency compensated signal to be used by the Lincompex demodulator.
Another example of the frequency compensation circuit 56 is a typical frequency shifting circuit or frequency translator as shown in FIG. 6(b). These circuits usually include oscillators, multipliers, and filters. In this example, (as shown in FIG. 6(b)), a multiplier 103 receives the incoming signal. The multiplier 103 is connected to a first frequency compensator oscillator 101 and a first filter 105. The first filter 105 is connected to a multiplier 109. Connected to multiplier 109 is a second frequency compensator oscillator and a second filter 111. The second frequency oscillator 107 is connected to the frequency measuring circuit 54. The second filter 111 is also connected to the input of the Lincompex demodulator.
A first frequency compensator oscillator 101 produces a frequency corresponding to a first frequency which is greater than the bandwidth of the signal being shifted, necessary to prevent the problem of overlap of the two side band signals which occur when the signal is multiplied by a frequency less than the bandwidth of the signal. For example, if the bandwidth of the combination information signal is 3000 Hz, it might be desirable to set the frequency F.sub.OSCA of the first frequency compensation oscillator 101 at 9000 Hz. The output of the first frequency compensation oscillator 101 is mixed with the combination information signal within a first frequency compensation mixer 103 and the lower sideband is filtered by a frequency compensation highpass filter 105 which removes the lower sideband produced by mixing the F.sub.OSCA with the combined information signal using mixer 103 to develop a frequency raised combination information signal. A second frequency compensation oscillator produces a frequency F.sub.OSCB which corresponds to the first frequency minus the frequency change. For example, if the combined information signal must be raised 12 Hz, the second frequency compensation oscillator 107 generates a frequency F.sub.OSC2 =F.sub.OSC1 -12, which is multiplied with the frequency raised combination information signal using a second frequency compensation mixer 109. This time a frequency compensation low pass filter 111 is used to remove the upper sideband, thereby raising (or lowering) the frequency of the combination information signal by the frequency error in this example, 12 Hz. In this prior art Lincompex System, the second frequency compensation oscillator 107 is a voltage controlled oscillator, thus producing a second frequency in accordance with the voltage level of the control signal received from the frequency measuring circuit 54.
With respect to the transmission of data, the Lincompex techniques used may or may not be different from the transmission of voice signals due to the complex nature of the data waveform. This technique is fully discussed in the '217 patent, and the frequency compensation technique therefor is fully discussed in the '451 patent. The frequency compensation technique, however, will be briefly described below.
As described in the '451 patent, a data signal is transmitted using Lincompex techniques. The problem associated with transmission of such data signal arises from the frequency band overlap occurring within such a complex waveform as shown in FIG. 2(b) where the data signal overlaps the envelope of the data signal. Furthermore, as described above with respect to the transmission of a voice signal, frequency drift can be experienced with the transmission of data signals.
FIGS. 4 and 5 illustrate collectively a Lincompex System where the input signal is frequency shifted prior to the compression/expansion operation to enable transmission of a wideband complex waveform using Lincompex techniques, and the received digital signal is frequency compensated to account for frequency drift. In FIGS. 4 and 5, as in all of the figures of the present invention, like elements throughout the drawing figures are identified with like numbers.
Whereas the FIGS. 1(a) and (b) prior art Lincompex systems normally receive speech or voice at their input 20, the embodiment of FIG. 4 would normally receive a data input such as a 16-tone parallel-tone or multi-tone data signal as illustrated in FIG. 7(a) at its input 20. However, it should be understood that any complex data waveform including voice may be transmitted over a communications channel.
A mixer 100 is provided to frequency shift the input data frequency spectra shown in FIG. 7(a), to a desired higher frequency F.sub.OSC to ensure that no overlap, as shown in FIG. 2(b), between the input data band, when compressed, and the control tone signal frequency band occurs. Accordingly, an oscillator 102 supplies the mixing frequency F.sub.OSC to a frequency multiplier or mixer 104 where it is mixed with the data input A of FIG. 7(a) to produce the mixed data B of FIG. 7(b). A filter 106 is then provided to band pass filter the mixed output to remove an undesired one of the two side bands produced by the mixing process. The frequency response of the filter 106 is illustrated in FIG. 7(c). After filtering, only a single side band is left which is frequency shifted to a desired frequency at the output D of the filter 106 as illustrated in FIG. 7(d). This frequency shifted data input of FIG. 7(d) is then treated by prior art Lincompex techniques by a compressor 22, envelope circuit 24, summer 32 and transmitter 34, in the manner described with respect to FIG. 1 to produce a combined data and envelope signal for transmission on the transmission medium 36. As explained previously, the compression operation enlarges the frequency spectrum of the compressed data as shown in FIG. 7(f). The envelope circuit 24 generates an envelope spectrum E which would have a selected bandwidth depending on various factors including the accuracy of compression to be achieved. However, it is important that the bandwidth of the envelope spectrum not overlap that of the compressed data so that full compression and recovery of the data can be accomplished by the Lincompex system.
FIG. 5 illustrates a prior art Lincompex demodulator which automatically frequency compensates the compressed complex waveform by frequency shifting the total bandwidth according to a measured frequency error prior to the splitting of the complex waveform into its components, a data signal and a control tone, thereby substantially eliminating data distortion due to frequency drift or the detuning of the transmitter/receiver system. The demodulator of FIG. 5, differs from that of FIG. 3 in that frequency shifted data is obtained from expander 42. This frequency shifted data may be provided via a line to an output where it may be detected by a detector sensitive to the frequency shifted tones contained within the frequency shifted data. Alternatively, as shown in FIG. 5, a receiver mixer 110 including a multiplier 108, frequency oscillator 112 and filter 114 may be utilized to shift the frequency shifted data back to its original frequency band. Accordingly, the digital information may be readily recovered.
While prior art Lincompex techniques allow elimination of frequency drift for a single channel frequency, the prior art Lincompex systems do no correct for the total frequency errors in cooperation with the frequency hopping techniques used by a telecommunications system.
Neither conventional frequency hopping techniques nor prior art Lincompex techniques eliminate frequency drift across a telecommunications link employing high frequency hopping rates.